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FEATURES High Speed 140 MHz Bandwidth (3 dB, G = +1) 120 MHz Bandwidth (3 dB, G = +2) 35 MHz Bandwidth (0.1 dB, G = +2) 2500 V/ s Slew Rate 25 ns Settling Time to 0.1% (For a 2 V Step) 65 ns Settling Time to 0.01% (For a 10 V Step) Excellent Video Performance (RL =150 ) 0.01% Differential Gain, 0.01 Differential Phase Voltage Noise of 1.9 nVHz Low Distortion: THD = -74 dB @ 10 MHz Excellent DC Precision 3 mV max Input Offset Voltage Flexible Operation Specified for 5 V and 15 V Operation 2.3 V Output Swing into a 75 Load (VS = 5 V) APPLICATIONS Video Crosspoint Switchers, Multimedia Broadcast Systems HDTV Compatible Systems Video Line Drivers, Distribution Amplifiers ADC/DAC Buffers DC Restoration Circuits Medical--Ultrasound, PET, Gamma and Counter Applications PRODUCT DESCRIPTION
NC 1 -IN 2 +IN 3 -VS 4 8 NC 7 +V S 6 OUTPUT
High Performance Video Op Amp AD811
NC NC NC 3 2 1 20 19 NC 4 NC 5 -IN 6 NC 7 +IN 8 9 10 11 12 13 -VS NC = NO CONNECT NC NC NC NC 18 NC NC NC
CONNECTION DIAGRAMS 20-Lead LCC (E-20A) Package 8-Lead Plastic (N-8) Cerdip (Q-8) SOIC (SO-8) Packages
AD811
17 NC 16 +V S 15 NC 14 OUTPUT
AD811
5 NC
NC = NO CONNECT
16-Lead SOIC (R-16) Package 20-Lead SOIC (R-20) Package
NC 1 NC 2 -IN 3 16 NC 15 NC 14 +V S 13 NC 12 OUTPUT 11 NC NC 1 20 NC 19 NC 18 NC 17 +V S 16 NC 15 OUTPUT 14 NC 13 NC
NC 2 NC 3 -IN 4 NC 5
NC 4 +IN NC 5 6
+IN 6 NC 7 -VS 8 NC 9 NC 10
-VS 7 NC 8
AD811
10 NC 9 NC
NC = NO CONNECT
AD811
12 NC 11 NC
NC = NO CONNECT
The AD811 is a wideband current-feedback operational amplifier, optimized for broadcast quality video systems. The -3 dB bandwidth of 120 MHz at a gain of +2 and differential gain and phase of 0.01% and 0.01 (RL = 150 ) make the AD811 an excellent choice for all video systems. The AD811 is designed to meet a stringent 0.1 dB gain flatness specification to a bandwidth of 35 MHz (G = +2) in addition to the low differential gain and phase errors. This performance is achieved whether driving one or two back terminated 75 cables, with a low power supply current of 16.5 mA. Furthermore, the AD811 is specified over a power supply range of 4.5 V to 18 V.
0.10 0.09 DIFFERENTIAL GAIN - % 0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01 5 6 7 8 9 10 11 12 13 14 GAIN PHASE RF = 649 FC = 3.58MHz 100 IRE MODULATED RAMP RL = 150 0.20 0.18 0.16 0.14 0.12 0.10 0.08 0.06 0.04 0.02 DIFFERENTIAL PHASE - Degrees
The AD811 is also excellent for pulsed applications where transient response is critical. It can achieve a maximum slew rate of greater than 2500 V/s with a settling time of less than 25 ns to 0.1% on a 2 volt step and 65 ns to 0.01% on a 10 volt step. The AD811 is ideal as an ADC or DAC buffer in data acquisition systems due to its low distortion up to 10 MHz and its wide unity gain bandwidth. Because the AD811 is a current feedback amplifier, this bandwidth can be maintained over a wide range of gains. The AD811 also offers low voltage and current noise of 1.9 nV/Hz and 20 pA/Hz, respectively, and excellent dc accuracy for wide dynamic range applications.
12 G = +2 RL = 150 RG = RFB 9 VS = 15V
6
GAIN - dB
3 VS = 0 -3 5V
-6 15 1M 10M FREQUENCY - Hz 100M SUPPLY VOLTAGE - Volts
REV. D
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 (c) Analog Devices, Inc., 1999
AD811-SPECIFICATIONS (@ T = +25 C and V =
A S
15 V dc, RLOAD = 150 unless otherwise noted)
Min AD811J/A1 Typ Max Min AD811S2 Typ Max Units
Model DYNAMIC PERFORMANCE Small Signal Bandwidth (No Peaking) -3 dB G = +1 G = +2 G = +2 G = +10 0.1 dB Flat G = +2 Full Power Bandwidth Slew Rate
3
Conditions
VS
RFB = 562 RFB = 649 RFB = 562 RFB = 511 RFB = 562 RFB = 649 VOUT = 20 V p-p VOUT = 4 V p-p VOUT = 20 V p-p 10 V Step, AV = -1 2 V Step, AV = -1 RFB = 649, AV = +2 f = 3.58 MHz f = 3.58 MHz VOUT = 2 V p-p, AV = +2 @ fC = 10 MHz
15 V 15 V 5 V 15 V 5 V 15 V 15 V 5 V 15 V 15 V 5 V 15 V 15 V 15 V 15 V 5 V 15 V 5 V, 15 V
140 120 80 100 25 35 40 400 2500 50 65 25 3.5 0.01 0.01 -74 36 43 0.5 5 3 5
140 120 80 100 25 35 40 400 2500 50 65 25 3.5 0.01 0.01 -74 36 43 0.5 5 5 15 10 20 2 2 5 30 10 25 3 5
MHz MHz MHz MHz MHz MHz MHz V/s V/s ns ns ns ns % Degree dBc dBm dBm mV mV V/C A A A A
Settling Time to 0.1% Settling Time to 0.01% Settling Time to 0.1% Rise Time, Fall Time Differential Gain Differential Phase THD @ fC = 10 MHz Third Order Intercept4 INPUT OFFSET VOLTAGE
TMIN to TMAX Offset Voltage Drift INPUT BIAS CURRENT -Input TMIN to TMAX +Input TMIN to TMAX TRANSRESISTANCE TMIN to TMAX VOUT = 10 V RL = RL = 200 VOUT = 2.5 V RL = 150 VCM = 2.5 VCM = 10 V TMIN to TMAX VS = 4.5 V to 18 V TMIN to TMAX TMIN to TMAX TMIN to TMAX f = 1 kHz f = 1 kHz 5 V 15 V TJ = +25C (Open Loop @ 5 MHz) 15 V 15 V 5 V 5 V 15 V 5 V, 15 V 5 V, 15 V
2 2
0.75 0.5 0.25
1.5 0.75 0.4
0.75 0.5
1.5 0.75
M M M
0.125 0.4
COMMON-MODE REJECTION VOS (vs. Common Mode) TMIN to TMAX TMIN to TMAX Input Current (vs. Common Mode) POWER SUPPLY REJECTION VOS +Input Current -Input Current INPUT VOLTAGE NOISE INPUT CURRENT NOISE OUTPUT CHARACTERISTICS Voltage Swing, Useful Operating Range5 Output Current Short-Circuit Current Output Resistance INPUT CHARACTERISTICS +Input Resistance -Input Resistance Input Capacitance Common-Mode Voltage Range POWER SUPPLY Operating Range Quiescent Current TRANSISTOR COUNT
56 60
60 66 1 70 0.3 0.4 1.9 20 2.9 12 100 150 9 1.5 14 7.5 3 13
50 56 3 60 2 2
60 66 1 70 0.3 0.4 1.9 20 2.9 12 100 150 9 1.5 14 7.5 3 13
3
dB dB A/V dB A/V A/V nV/Hz pA/Hz V V mA mA M pF V V
60
2 2
+Input
5 V 15 V 5 V 15 V 4.5
14.5 16.5 40
18 16.0 18.0
4.5 14.5 16.5 40
18 16.0 18.0
V mA mA
# of Transistors
NOTES 1 The AD811JR is specified with 5 V power supplies only, with operation up to 12 volts. 2 See Analog Devices' military data sheet for 883B tested specifications. 3 FPBW = slew rate/(2 VPEAK). 4 Output power level, tested at a closed loop gain of two. 5 Useful operating range is defined as the output voltage at which linearity begins to degrade. Specifications subject to change without notice.
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REV. D
AD811
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V AD811JR Grade Only . . . . . . . . . . . . . . . . . . . . . . . . . 12 V Internal Power Dissipation2 . . . . . . . . Observe Derating Curves Output Short Circuit Duration . . . . . Observe Derating Curves Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . VS Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . 6 V Storage Temperature Range (Q, E) . . . . . . . . -65C to +150C Storage Temperature Range (N, R) . . . . . . . . -65C to +125C Operating Temperature Range AD811J . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to +70C AD811A . . . . . . . . . . . . . . . . . . . . . . . . . . . . -40C to +85C AD811S . . . . . . . . . . . . . . . . . . . . . . . . . . . -55C to +125C Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300C
NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 8-Lead Plastic Package: JA = 90C/W 8-Lead Cerdip Package: JA = 110C/W 8-Lead SOIC Package: JA = 155C/W 16-Lead SOIC Package: JA = 85C/W 20-Lead SOIC Package: JA = 80C/W 20-Lead LCC Package: JA = 70C/W
ABSOLUTE MAXIMUM RATINGS 1
MAXIMUM POWER DISSIPATION
The maximum power that can be safely dissipated by the AD811 is limited by the associated rise in junction temperature. For the plastic packages, the maximum safe junction temperature is +145C. For the cerdip and LCC packages, the maximum junction temperature is +175C. If these maximums are exceeded momentarily, proper circuit operation will be restored as soon as the die temperature is reduced. Leaving the device in the "overheated" condition for an extended period can result in device burnout. To ensure proper operation, it is important to observe the derating curves in Figures 17 and 18. While the AD811 is internally short circuit protected, this may not be sufficient to guarantee that the maximum junction temperature is not exceeded under all conditions. One important example is when the amplifier is driving a reverse terminated 75 cable and the cable's far end is shorted to a power supply. With power supplies of 12 volts (or less) at an ambient temperature of +25C or less, if the cable is shorted to a supply rail, then the amplifier will not be destroyed, even if this condition persists for an extended period.
ESD SUSCEPTIBILITY
ORDERING GUIDE
Model AD811AN AD811AR-16 AD811AR-20 AD811JR AD811SQ/883B 5962-9313101MPA AD811SE/883B 5962-9313101M2A AD811JR-REEL AD811JR-REEL7 AD811AR-16-REEL AD811AR-16-REEL7 AD811AR-20-REEL AD811ACHIPS AD811SCHIPS
Temperature Range -40C to +85C -40C to +85C -40C to +85C 0C to +70C -55C to +125C -55C to +125C -55C to +125C -55C to +125C 0C to +70C 0C to +70C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -55C to +125C
Package Option* N-8 R-16 R-20 SO-8 Q-8 Q-8 E-20A E-20A SO-8 SO-8 R-16 R-16 R-20 Die Die
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 volts, which readily accumulate on the human body and on test equipment, can discharge without detection. Although the AD811 features proprietary ESD protection circuitry, permanent damage may still occur on these devices if they are subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid any performance degradation or loss of functionality.
METALIZATION PHOTOGRAPH
Contact Factory for Latest Dimensions. Dimensions Shown in Inches and (mm).
*E = Ceramic Leadless Chip Carrier; N = Plastic DIP; Q = Cerdip; SO (R) = Small Outline IC (SOIC).
CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD811 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
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AD811-Typical Performance Characteristics
Volts
TA = +25 C 15
Volts
20
20 TA = +25 C 15
MAGNITUDE OF THE OUTPUT VOLTAGE -
COMMON-MODE VOLTAGE RANGE -
10
NO LOAD 10
5
RL = 150 5
0 0 5 10 SUPPLY VOLTAGE - Volts 15 20
0 0 5 10 SUPPLY VOLTAGE - Volts 15 20
Figure 1. Input Common-Mode Voltage Range vs. Supply
Figure 4. Output Voltage Swing vs. Supply
35
21
QUIESCENT SUPPLY CURRENT - mA
30
OUTPUT VOLTAGE - Volts p-p
18
VS = 25 20 15
15V
15
VS =
15V
12 VS = 9 5V
VS = 10 5 0 10
5V
6
1k 100 LOAD RESISTANCE -
10k
3 -60
-40
-20
0 20 60 40 80 100 JUNCTION TEMPERATURE - C
120
140
Figure 2. Output Voltage Swing vs. Resistive Load
Figure 5. Quiescent Supply Current vs. Junction Temperature
10 NONINVERTING INPUT 5 TO 15V
INPUT OFFSET VOLTAGE - mV
10 8 6 4 2 0 -2 -4 -6 -8 VS = 15V VS = 5V
A INPUT BIAS CURRENT -
0 VS = INVERTING INPUT -10 5V
-20
VS =
15V
-30 -60
-40
-20
0 20 40 60 80 JUNCTION TEMPERATURE - C
100
120
140
-10 -60
-40
-20
0
20
40
60
80
100
120
140
JUNCTION TEMPERATURE - C
Figure 3. Input Bias Current vs. Junction Temperature
Figure 6. Input Offset Voltage vs. Junction Temperature
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REV. D
AD811
250
2.0 VS = 15V RL = 200 VOUT = 10V
TRANSRESISTANCE - M
SHORT CIRCUIT CURRENT - mA
200 VS = 150 15V
1.5
1.0
100
VS =
5V
0.5
VS = 5V RL = 150 VOUT = 2.5V
50 -60
-40
-20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
0 -60 -40
-20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
Figure 7. Short Circuit Current vs. Junction Temperature
Figure 10. Transresistance vs. Junction Temperature
10 CLOSED-LOOP OUTPUT RESISTANCE -
100
100
1
NOISE VOLTAGE - nV/ Hz
10
INVERTING CURRENT VS =
5 TO 15V 10
0.1 VS = 15V GAIN = +2 RFB = 649 0.01 10k
VOLTAGE NOISE VS =
15V
VOLTAGE NOISE VS = 1
5V 10k 1 100k
100k
1M FREQUENCY - Hz
10M
100M
10
100
1k FREQUENCY - Hz
Figure 8. Closed-Loop Output Resistance vs. Frequency
Figure 11. Input Noise vs. Frequency
10 RISE TIME 8 60
-3dB BANDWIDTH - MHz
OVERSHOOT - %
200 VO = 1V p-p VS= 15V RL= 150 GAIN = +2
10
160
8
RISETIME - ns
6 OVERSHOOT 4 VS = 15V VO = 1V p-p RL = 150 GAIN = +2 2
40
120 BANDWIDTH 80
6
20
4
0
40 PEAKING
2
0 400
1.0k 1.2k 1.4k 800 VALUE OF FEEDBACK RESISTOR (RFB) -
600
1.6k
0 400
600
1.0k 1.2k 1.4k 800 VALUE OF FEEDBACK RESISTOR (RFB) -
0 1.6k
Figure 9. Rise Time and Overshoot vs. Value of Feedback Resistor, RFB
Figure 12. 3 dB Bandwidth and Peaking vs. Value of RFB
REV. D
-5-
PEAKING - dB
NOISE CURRENT - pA/ Hz
VS =
5V
NONINVERTING CURRENT VS =
5 TO 15V
AD811
110 649 100 VIN 90 150 80 70 VS = 60 50 40 30 1k 10k 100k FREQUENCY - Hz 1M 10M VS = 5V 15V 150 649 VOUT
25
OUTPUT VOLTAGE - Volts p-p
20
VS =
15V
CMRR - dB
15
GAIN = +10 OUTPUT LEVEL FOR 3% THD
10 VS = 5V
5
0 100k
1M 10M FREQUENCY - Hz
100M
Figure 13. Common-Mode Rejection vs. Frequency
Figure 16. Large Signal Frequency Response
80 70 60 50 40 30 20 10 5 1k 10k 100k FREQUENCY - Hz 1M 10M
CURVES ARE FOR WORST CASE CONDITION WHERE ONE SUPPLY IS VARIED WHILE THE OTHER IS HELD CONSTANT.
-50 VS = 15V RF = 649 AV = +2 VOUT = 2V p-p RL = 100 GAIN = +2 -70 5V SUPPLIES
VS =
5V
HARMONIC DISTORTION - dBc
PSRR - dB
-90
2ND HARMONIC 3RD HARMONIC
15V SUPPLIES
-110
2ND HARMONIC 3RD HARMONIC
-130 1k 10k 100k FREQUENCY - Hz 1M 10M
Figure 14. Power Supply Rejection vs. Frequency
Figure 17. Harmonic Distortion vs. Frequency
2.5 TJ MAX = +145 C
3.4 3.2 TOTAL POWER DISSIPATION - Watts 3.0 2.8 2.6 2.4 2.2 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 -60 8-LEAD CERDIP 20-LEAD LCC TJ MAX = +175 C
TOTAL POWER DISSIPATION - Watts
16-LEAD SOIC 2.0 20-LEAD SOIC 1.5 8-LEAD MINI-DIP
1.0 8-LEAD SOIC 0.5 -50 -40 -30 -20 -10 0 10 20 30 40 50 AMBIENT TEMPERATURE - C
60 70
80
90
-40
-20
0 20 40 60 80 AMBIENT TEMPERATURE - C
100
120
140
Figure 15. Maximum Power Dissipation vs. Temperature for Plastic Packages
Figure 18. Maximum Power Dissipation vs. Temperature for Hermetic Packages
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REV. D
Typical Characteristics, Noninverting Connection-AD811
9
RFB +VS 0.1 F RG VOUT TO TEKTRONIX P6201 FET PROBE RL
6 3
G = +1 RL = 150 RG = VS = 15V RFB = 750
GAIN - dB
0 -3 VS = 5V RFB = 619
VIN HP8130 50 PULSE GENERATOR
AD811
+
-6
-VS
0.1 F
-9 -12 1M 10M FREQUENCY - Hz 100M
Figure 19. Noninverting Amplifier Connection
Figure 22. Closed-Loop Gain vs. Frequency, Gain = +1
26
1V
100
10ns
23
G = +10 RL = 150
VS = 15V RFB = 511
VIN
90
20
GAIN - dB
17 VS = 5V R FB = 442
14
VOUT 10
0%
11
1V
8 1M 10M FREQUENCY - Hz 100M
Figure 20. Small Signal Pulse Response, Gain = +1
Figure 23. Closed-Loop Gain vs. Frequency, Gain = +10
100mV
100
10ns
100
1V
VIN
90
20ns
VIN
90
VOUT 10
0%
VOUT 10
0%
1V
10V
Figure 21. Small Signal Pulse Response, Gain = +10
Figure 24. Large Signal Pulse Response, Gain = +10
REV. D
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AD811-Typical Characteristics, Inverting Connection
RFB +VS 0.1 F VIN HP8130 PULSE GENERATOR RG VOUT TO TEKTRONIX P6201 FET PROBE 6 G = -1 RL = 150 VS = 15V RFB = 590 3
0
GAIN - dB
-3 V S = 5V RFB = 562
AD811
RL
-6
-9 0.1 F -12 1M -VS 10M FREQUENCY - Hz 100M
Figure 25. Inverting Amplifier Connection
Figure 28. Closed-Loop Gain vs. Frequency, Gain = -1
26
1V
100
10ns
23
G = -10 RL = 150
VS = 15V RFB = 511
VIN
90
20
GAIN - dB
17
14
VOUT 10
0%
VS = 5V RFB = 442
11
1V
8 1M 10M FREQUENCY - Hz 100M
Figure 26. Small Signal Pulse Response, Gain = -1
Figure 29. Closed-Loop Gain vs. Frequency, Gain = -10
100mV
100
10ns
100
1V
VIN
90
20ns
VIN
90
VOUT 10
0%
VOUT 10
0%
1V
Figure 27. Small Signal Pulse Response, Gain = -10
10V
Figure 30. Large Signal Pulse Response, Gain = -10
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REV. D
AD811
APPLICATIONS General Design Considerations Achieving the Flattest Gain Response at High Frequency
The AD811 is a current feedback amplifier optimized for use in high performance video and data acquisition applications. Since it uses a current feedback architecture, its closed-loop -3 dB bandwidth is dependent on the magnitude of the feedback resistor. The desired closed-loop gain and bandwidth are obtained by varying the feedback resistor (RFB) to tune the bandwidth, and varying the gain resistor (RG) to get the correct gain. Table I contains recommended resistor values for a variety of useful closed-loop gains and supply voltages.
Table I. -3 dB Bandwidth vs. Closed-Loop Gain and Resistance Values
Achieving and maintaining gain flatness of better than 0.1 dB at frequencies above 10 MHz requires careful consideration of several issues.
Choice of Feedback and Gain Resistors
VS = 15 V Closed-Loop Gain +1 +2 +10 -1 -10 VS = 5 V Closed-Loop Gain +1 +2 +10 -1 -10 VS = 10 V Closed-Loop Gain +1 +2 +10 -1 -10
Because of the above-mentioned relationship between the 3 dB bandwidth and the feedback resistor, the fine scale gain flatness will, to some extent, vary with feedback resistor tolerance. It is, therefore, recommended that resistors with a 1% tolerance be used if it is desired to maintain flatness over a wide range of production lots. In addition, resistors of different construction have different associated parasitic capacitance and inductance. Metal-film resistors were used for the bulk of the characterization for this data sheet. It is possible that values other than those indicated will be optimal for other resistor types.
Printed Circuit Board Layout Considerations
RFB 750 649 511 590 511
RG 649 56.2 590 51.1
-3 dB BW (MHz) 140 120 100 115 95 -3 dB BW (MHz) 80 80 65 75 65 -3 dB BW (MHz) 105 105 80 105 80
RFB 619 562 442 562 442
RG 562 48.7 562 44.2
As to be expected for a wideband amplifier, PC board parasitics can affect the overall closed loop performance. Of concern are stray capacitances at the output and the inverting input nodes. If a ground plane is to be used on the same side of the board as the signal traces, a space (3/16" is plenty) should be left around the signal lines to minimize coupling. Additionally, signal lines connecting the feedback and gain resistors should be short enough so that their associated inductance does not cause high frequency gain errors. Line lengths less than 1/4" are recommended.
Quality of Coaxial Cable
Optimum flatness when driving a coax cable is possible only when the driven cable is terminated at each end with a resistor matching its characteristic impedance. If the coax was ideal, then the resulting flatness would not be affected by the length of the cable. While outstanding results can be achieved using inexpensive cables, it should be noted that some variation in flatness due to varying cable lengths may be experienced.
Power Supply Bypassing
RFB 649 590 499 590 499
RG 590 49.9 590 49.9
Figures 11 and 12 illustrate the relationship between the feedback resistor and the frequency and time domain response characteristics for a closed-loop gain of +2. (The response at other gains will be similar.) The 3 dB bandwidth is somewhat dependent on the power supply voltage. As the supply voltage is decreased for example, the magnitude of internal junction capacitances is increased, causing a reduction in closed-loop bandwidth. To compensate for this, smaller values of feedback resistor are used at lower supply voltages.
Adequate power supply bypassing can be critical when optimizing the performance of a high frequency circuit. Inductance in the power supply leads can form resonant circuits that produce peaking in the amplifier's response. In addition, if large current transients must be delivered to the load, then bypass capacitors (typically greater than 1 F) will be required to provide the best settling time and lowest distortion. Although the recommended 0.1 F power supply bypass capacitors will be sufficient in many applications, more elaborate bypassing (such as using two paralleled capacitors) may be required in some cases.
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AD811
Driving Capacitive Loads
100 90 80 70
VALUE OF RS -
The feedback and gain resistor values in Table I will result in very flat closed-loop responses in applications where the load capacitances are below 10 pF. Capacitances greater than this will result in increased peaking and overshoot, although not necessarily in a sustained oscillation. There are at least two very effective ways to compensate for this effect. One way is to increase the magnitude of the feedback resistor, which lowers the 3 dB frequency. The other method is to include a small resistor in series with the output of the amplifier to isolate it from the load capacitance. The results of these two techniques are illustrated in Figure 32. Using a 1.5 k feedback resistor, the output ripple is less than 0.5 dB when driving 100 pF. The main disadvantage of this method is that it sacrifices a little bit of gain flatness for increased capacitive load drive capability. With the second method, using a series resistor, the loss of flatness does not occur.
RFB +VS 0.1 F RG RS (OPTIONAL) VIN
G = +2 VS = 15V RS VALUE SPECIFIED IS FOR FLATTEST FREQUENCY RESPONSE
60 50 40 30 20 10 0 10 100 LOAD CAPACITANCE - pF 1000
Figure 33. Recommended Value of Series Resistor vs. the Amount of Capacitive Load
Figure 33 shows recommended resistor values for different load capacitances. Refer again to Figure 32 for an example of the results of this method. Note that it may be necessary to adjust the gain setting resistor, RG, to correct for the attenuation which results due to the divider formed by the series resistor, RS, and the load resistance.
VOUT CL RL
AD811
RT 0.1 F
Applications which require driving a large load capacitance at a high slew rate are often limited by the output current available from the driving amplifier. For example, an amplifier limited to 25 mA output current cannot drive a 500 pF load at a slew rate greater than 50 V/s. However, because of the AD811's 100 mA output current, a slew rate of 200 V/s is achievable when driving this same 500 pF capacitor (see Figure 34).
2V
100
-VS
Figure 31. Recommended Connection for Driving a Large Capacitive Load
12
100ns
VIN
90
9
RFB = 1.5k RS = 0
6
GAIN - dB
VOUT 10
3 G = +2 VS = 15V RL = 10k CL = 100pF
0%
RFB = 649 RS = 30
0
5V
Figure 34. Output Waveform of an AD811 Driving a 500 pF Load. Gain = +2, RFB = 649 , RS = 15 , RS = 10 k
-3
-6 1M 10M FREQUENCY - Hz 100M
Figure 32. Performance Comparison of Two Methods for Driving a Capacitive Load
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REV. D
AD811
Operation as a Video Line Driver
The AD811 has been designed to offer outstanding performance at closed-loop gains of one or greater, while driving multiple reverse-terminated video loads. The lowest differential gain and phase errors will be obtained when using 15 volt power supplies. With 12 volt supplies, there will be an insignificant increase in these errors and a slight improvement in gain flatness. Due to power dissipation considerations, 12 volt supplies are recommended for optimum video performance. Excellent performance can be achieved at much lower supplies as well. The closed-loop gain vs. frequency at different supply voltages is shown in Figure 36. Figure 37 is an oscilloscope photograph of an AD811 line driver's pulse response with 15 volt supplies. The differential gain and phase error vs. supply are plotted in Figures 38 and 39, respectively. Another important consideration when driving multiple cables is the high frequency isolation between the outputs of the cables. Due to its low output impedance, the AD811 achieves better than 40 dB of output to output isolation at 5 MHz driving back terminated 75 cables.
649 649 75 +VS 0.1 F 75 75 CABLE VOUT #1
1V
100
10ns
VIN
90
VOUT 10
0%
1V
Figure 37. Small Signal Pulse Response, Gain = +2, VS = 15 V
0.10 0.09
DIFFERENTIAL GAIN - %
0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01
RF= 649 FC= 3.58MHz 100 IRE MODULATED RAMP
a. DRIVING A SINGLE, BACK TERMINATED,
75 COAX CABLE
b. DRIVING TWO PARALLEL,
BACK TERMINATED, COAX CABLES
75 VIN 75 CABLE
CABLE VOUT #2
AD811
b
75 75
a
75 0.1 F
5
6
7
8 9 10 11 12 SUPPLY VOLTAGE - Volts
13
14
15
-VS
Figure 38. Differential Gain Error vs. Supply Voltage for the Video Line Driver of Figure 35
Figure 35. A Video Line Driver Operating at a Gain of +2
12 G = +2 RL = 150 RG = RFB VS = 15V RFB = 649
0.20
DIFFERENTIAL PHASE - Degrees
0.18 0.16 0.14 0.12 0.10 0.08 0.06 0.04
9
RF = 649 FC = 3.58MHz 100 IRE MODULATED RAMP
6
a. DRIVING A SINGLE, BACK TERMINATED,
75 COAX CABLE
GAIN - dB
3 VS = 5V RFB = 562 0
b. DRIVING TWO PARALLEL,
BACK TERMINATED, COAX CABLES
b
-3
0.02
-6 1 10 FREQUENCY - MHz 100
a
5 6 7 8 9 10 11 12 SUPPLY VOLTAGE - Volts 13 14 15
Figure 36. Closed-Loop Gain vs. Frequency, Gain = +2
Figure 39. Differential Phase Error vs. Supply Voltage for the Video Line Driver of Figure 35
REV. D
-11-
AD811
An 80 MHz Voltage-Controlled Amplifier Circuit
The voltage-controlled amplifier (VCA) circuit of Figure 40 shows the AD811 being used with the AD834, a 500 MHz, 4-quadrant multiplier. The AD834 multiplies the signal input by the dc control voltage, VG. The AD834 outputs are in the form of differential currents from a pair of open collectors, ensuring that the full bandwidth of the multiplier (which exceeds 500 MHz) is available for certain applications. Here, the AD811 op amp provides a buffered, single-ended groundreferenced output. Using feedback resistors R8 and R9 of 511 , the overall gain ranges from -70 dB, for VG = 0 dB to +12 dB, (a numerical gain of four), when VG = +1 V. The overall transfer function of the VCA is: VOUT = 4 (X1 - X2)(Y1 - Y2) which reduces to VOUT = 4 VG VIN using the labeling conventions shown in Figure 40. The circuit's -3 dB bandwidth of 80 MHz, is maintained essentially constant--independent of gain. The response can be maintained flat to within 0.1 dB from dc to 40 MHz at full gain with the addition of an optional capacitor of about 0.3 pF across the feedback resistor R8. The circuit produces a full-scale output of 4 V for a 1 V input, and can drive a reverse-terminated load of 50 or 75 to 2 V.
The gain can be increased to 20 dB (x10) by raising R8 and R9 to 1.27 k, with a corresponding decrease in -3 dB bandwidth to about 25 MHz. The maximum output voltage under these conditions will be increased to 9 V using 12 V supplies. The gain-control input voltage, VG, may be a positive or negative ground-referenced voltage, or fully differential, depending on the user's choice of connections at Pins 7 and 8. A positive value of VG results in an overall noninverting response. Reversing the sign of VG simply causes the sign of the overall response to invert. In fact, although this circuit has been classified as a voltage-controlled amplifier, it is also quite useful as a generalpurpose four-quadrant multiplier, with good load-driving capabilities and fully-symmetrical responses from X- and Y-inputs. The AD811 and AD834 can both be operated from power supply voltages of 5 V. While it is not necessary to power them from the same supplies, the common-mode voltage at W1 and W2 must be biased within the common-mode range of the AD811's input stage. To achieve the lowest differential gain and phase errors, it is recommended that the AD811 be operated from power supply voltages of 10 volts or greater. This VCA circuit is designed to operate from a 12 volt dual power supply.
FB +12V C1 0.1 F
+
VG
R1 100
R8*
-
R2 100
8 X2
7
6
5 W1
X1 +V S
R4 182
R6 294
U1 AD834
U3 AD811
W2 4
VOUT
Y1 1 VIN
Y2 2
-VS 3
R5 182
R7 294
RL
R3 249
R9* C2 0.1 F
-12V *R8 = R9 = 511 FOR X4 GAIN = 1.27k FOR X10 GAIN FB
Figure 40. An 80 MHz Voltage-Controlled Amplifier
-12-
REV. D
AD811
A Video Keyer Circuit
By using two AD834 multipliers, an AD811, and a 1 V dc source, a special form of a two-input VCA circuit called a video keyer can be assembled. "Keying" is the term used in reference to blending two or more video sources under the control of a third signal or signals to create such special effects as dissolves and overlays. The circuit shown in Figure 41 is a two-input keyer, with video inputs VA and VB, and a control input VG. The transfer function (with VOUT at the load) is given by: VOUT = G VA + (1-G) VB where G is a dimensionless variable (actually, just the gain of the "A" signal path) that ranges from 0 when VG = 0, to 1 when VG = +1 V. Thus, VOUT varies continuously between VA and VB as G varies from 0 to 1. Circuit operation is straightforward. Consider first the signal path through U1, which handles video input VA. Its gain is clearly zero when VG = 0 and the scaling we have chosen ensures that it is unity when VG = +1 V; this takes care of the first term of the transfer function. On the other hand, the VG input to U2 is taken to the inverting input X2 while X1 is biased at an accurate +1 V. Thus, when VG = 0, the response to video input VB is already at its full-scale value of unity, whereas when VG = +1 V, the differential input X1-X2 is zero. This generates the second term.
+5V R7 45.3 R6 226 8 X2 +5V R1 1.87k U4 AD589 Y1 1 7 6 X1 +VS 5 W1
The bias currents required at the output of the multipliers are provided by R8 and R9. A dc-level-shifting network comprising R10/R12 and R11/R13 ensures that the input nodes of the AD811 are positioned at a voltage within its common-mode range. At high frequencies C1 and C2 bypass R10 and R11 respectively. R14 is included to lower the HF loop gain, and is needed because the voltage-to-current conversion in the AD834s, via the Y2 inputs, results in an effective value of the feedback resistance of 250 ; this is only about half the value required for optimum flatness in the AD811's response. (Note that this resistance is unaffected by G: when G = 1, all the feedback is via U1, while when G = 0 it is all via U2). R14 reduces the fractional amount of output current from the multipliers into the current-summing inverting input of the AD811, by sharing it with R8. This resistor can be used to adjust the bandwidth and damping factor to best suit the application. To generate the 1 V dc needed for the "1-G" term an AD589 reference supplies 1.225 V 25 mV to a voltage divider consisting of resistors R2 through R4. Potentiometer R3 should be adjusted to provide exactly +1 V at the X1 input. In this case, we have shown an arrangement using dual supplies of 5 V for both the AD834 and the AD811. Also, the overall gain in this case is arranged to be unity at the load, when it is driven from a reverse-terminated 75 line. This means that the "dual VCA" has to operate at a maximum gain of 2, rather
C1 0.1 F
R14 SEE TEXT
SETUP FOR DRIVING REVERSE-TERMINATED LOAD TO PIN 6 AD811 ZO 200 TO Y2 200 VOUT ZO
R5 113 VG (0 TO +1V dc)
R10 2.49k
U1 AD834
Y2 2 -VS 3 W2 4
INSET
R8 29.4 R12 6.98k +5V
R2 174 VA ( 1V FS) R3 100
FB -5V +5V -5V C3 0.1 F LOAD GND
R4 1.02k
8 X2
7
6
5 W1
X1 +V S
R9 29.4
R13 6.98k
U3 AD811
C4 0.1 F
VOUT
U1 AD834
C2 0.1 F
Y1 1 VB ( 1V FS)
Y2 2
-VS 3 -5V
W2 4 R11 2.49k
FB
LOAD GND
-5V
Figure 41. A Practical Video Keyer Circuit
REV. D
-13-
AD811
than 4 as in the VCA circuit of Figure 40. However, this cannot be achieved by lowering the feedback resistor, since below a critical value (not much less than 500 ) the AD811's peaking may be unacceptable. This is because the dominant pole in the open-loop ac response of a current-feedback amplifier is controlled by this feedback resistor. It would be possible to operate at a gain of X4 and then attenuate the signal at the output. Instead, we have chosen to attenuate the signals by 6 dB at the input to the AD811; this is the function of R8 through R11. Figure 42 is a plot of the ac response of the feedback keyer, when driving a reverse terminated 50 cable. Output noise and adjacent channel feedthrough, with either channel fully off and the other fully on, is about -50 dB to 10 MHz. The feedthrough at 100 MHz is limited primarily by board layout. For VG = +1 V, the -3 dB bandwidth is 15 MHz when using a 137 resistor for R14 and 70 MHz with R14 = 49.9 . For further information regarding the design and operation of the VCA and video keyer circuits, refer to the application note "Video VCA's and Keyers Using the AD834 & AD811" by Brunner, Clarke, and Gilbert, available FREE from Analog Devices.
R14 = 49.9 0 -10
CLOSED-LOOP GAIN - dB
GAIN R14 = 137
-20 -30 -40 -50 -60 -70 -80 ADJACENT CHANNEL FEEDTHROUGH
10k
100k
1M FREQUENCY - Hz
10M
100M
Figure 42. A Plot of the AC Response of the Video Keyer
-14-
REV. D
AD811
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8- Lead Plastic DIP (N) Package
0.39 (9.91) MAX
8 5
20-Lead LCC (E-20A) Package
0.082 0.018 (2.085 0.455) 0.350 0.008 SQ (8.89 0.20) SQ
0.25 0.31 (6.35) (7.87)
1 4
PIN 1 0.10 (2.54) BSC 0.165 0.01 (4.19 0.25) 0.125 (3.18) MIN 0.018 (0.46 0.003 0.08) 0.033 (0.84) NOM 0.035 (0.89 0.01 0.25) 0.18 0.03 (4.57 0.75) SEATING PLANE
0.025 0.003 (0.635 0.075)
0.30 (7.62) REF
0.050 (1.27)
NO. 1 PIN INDEX
0.011 (0.28 15 0
0.003 0.08)
0.020 x 45 (0.51 x 45) REF
16-Lead SOIC (R-16) Package
8-Lead Cerdip (Q) Package
16 9 0.299 (7.60) 0.291 (7.40) 0.419 (10.65) 0.404 (10.26)
0.005 (0.13) MIN
8
0.055 (1.4) MAX
5
PIN 1
1 4
0.310 (7.87) 0.220 (5.59)
PIN 1 1 8
0.100 (2.54) BSC 0.405 (10.29) MAX 0.200.(5.08) MAX 0.200 (5.08) 0.125 (3.18) 0.060 (1.52) 0.015 (0.38) 0.150 (3.81) MIN 15 0 0.015 (0.38) 0.008 (0.20) 0.320 (8.13) 0.290 (7.37)
0.413 (10.50) 0.398 (10.10)
0.107 (2.72) 0.089 (2.26)
0.364 (9.246) 0.344 (8.738)
0.010 (0.25) 0.004 (0.10)
0.050 (1.27) BSC
0.018 (0.46) 0.014 (0.36)
SEATING 0.023 (0.58) 0.070 (1.78) PLANE 0.014 (0.36) 0.030 (0.76)
0.015 (0.38) 0.007 (1.18)
0.045 (1.15) 0.020 (0.50)
20-Lead Wide Body SOIC (R-20) Package 8-Lead SOIC (SO-8) Package
0.512 (13.00) 0.496 (12.60)
0.1968 (5.00) 0.1890 (4.80)
20 11
8
5 4
0.1574 (4.00) 0.1497 (3.80) PIN 1
1
0.2440 (6.20) 0.2284 (5.80)
0.300 (7.60) 0.292 (7.40) 0.419 (10.65) 0.394 (10.00)
0.0098 (0.25) 0.0040 (0.10) SEATING PLANE
0.0688 (1.75) 0.0532 (1.35) 0.0192 (0.49) 0.0138 (0.35) 8 0.0098 (0.25) 0 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41)
0.50 (1.27) BSC 0.019 (0.48) 0.014 (0.36) 0.104 (2.64) 0.093 (2.36) 0.011 (0.28) 0.004 (0.10) 0.015 (0.38) 0.007 (0.18) 0.050 (1.27) 0.016 (0.40)
REV. D
-15-
PRINTED IN U.S.A.
0.0500 (1.27) BSC
0.0196 (0.50) 0.0099 (0.25)
45
1
10
C1592b-0-8/99
0.040 x 45 (1.02 x 45) REF 3 PLCS

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